Typically, in electromagnetic flow meters for measuring the flow rate of a fluid that is electrically conductive, the flow rate of the fluid that flows within a pipe is measured by providing a magnetic excitation electric current that alternatingly switches polarities to a magnetic excitation coil that is disposed so that the direction of the magnetic field that is produced is perpendicular to the direction of flow of the fluid that is flowing within the pipe, to detect and differentially amplify, using a signal amplifying circuit, the electromotive force that is produced between a pair of detector electrodes that are disposed within the pipe perpendicular to the magnetic field produced by the magnetic excitation coil, and sampling and performing signal processing on the flow rate signal that is produced. See, for example, Japanese Unexamined Patent Application Publication No. H08-021756.
At this time, having the magnetic excitation current be as large as possible produces a large flow rate signal, which can improve the measurement accuracy. However, because the ramp up of the magnetic excitation electric current is slow because of the inductance in the magnetic excitation coil, if the flow rate signal is to be sampled in a steady domain, the larger the electric current, the lower the magnetic excitation frequency must be. On the other hand, when the magnetic excitation frequency is reduced, the fluid noise (noise that has the property of being inversely proportional to the frequency) increases, and has an adverse effect on the signal-to-noise ratio. Because of this, in typical electromagnetic flow meters, a magnetic excitation frequency of between about ⅛ and ¼ of the commercial power supply frequency is used.
Moreover, in the signal amplifying circuit, the input impedance of the signal amplifying circuit must be as high as possible so as to not produce attenuation in the flow rate signal, even if the fluid that is being measured has low electrical resistivity. Moreover, so that the contact resistance of the detecting electrodes does not increase due to an insulating substance is adhering to the detecting electrodes through the occurrence of electrochemical reactions between the detecting electrodes in the pipe and the fluid interface, the DC electric current that flows from the signal amplifying circuit to the detecting electrode side, and the DC electric current that flows from the detecting electrodes into the signal amplifying circuit, must be as small as possible.
FIG. 6 is a conventional signal amplifying circuit. Here, in a signal amplifying circuit 52 in a converter 50, the input impedance is increased through the preamplifiers U11 and U12 at the first stage. Moreover, the adherence of insulating substances through “electrochemical reactions between the detecting electrodes of the detector and the fluid interface,” as mentioned above, are suppressed through the use of FET input-type operational amplifiers, with input bias currents that are as small as possible, for U11 and U12.
Moreover, although the input impedance of the FET-input operational amplifier is extremely high, and the input bias current is also extremely low, to that extent it becomes susceptible to the effects of noise produced by the peripheral circuitry, and the like. Moreover, when the insulating characteristics of the circuit board degrade due to the effects of ambient moisture or adhesion of contaminating particles, the input impedance will be reduced.
Because of this, the interconnection patterns L11 and L12 from the connector CN11 for connecting the signal cables to the non-inverting input terminals of U11 and U12 are surrounded by guard rings GR11 and GR12 by the interconnection patterns for the output terminals of U11 and U12 (which are at the same electropotential as the input signals and which have low impedance), so as to prevent adverse effects of the peripheral circuitry and to prevent adverse effects of any degradation in the insulating properties of the circuit board.
FIG. 7 is a block diagram illustrating a configuration for a conventional electromagnetic flow meter. When a detector 60 is of a discrete type and is disposed at a location that is away from the converter 50, the detector 60 and the converter 50 are connected by a double-shielded line 61, as illustrated in FIG. 7. In this case, attenuation of the flow rate signal due to the capacitance between lines is prevented through applying the same voltage as the guard ring in FIG. 6, described above, to the inside shield lines SA and SB.
Moreover, the outputs of U11 and U12 in FIG. 6 are connected also to a fault detecting circuit 54 of FIG. 7, where if there ceases to be fluid within the pipe Pex so that the detecting electrodes TA and TB of the detector 60 cease to contact the fluid, or if a signal line in the detector 60 breaks or becomes disconnected, or the like, or if the output levels of U11 and U12 otherwise go outside of the normal range, the controlling circuit 56 is notified that there is a fault in the input signal. In response, the controlling circuit 56 performs fault handling through turning ON an alarm output from a digital output circuit 58, and sets the output level of an analog output circuit 57 to a level for when a fault is detected.
In such conventional technology, an input offset voltage for the buffer amplifiers U11 and U12 has thermal characteristics (offset voltage temperature drift) of about several mV/1° C., even in the high precision type, where this DC component is outputted from U11 and U12. These thermal characteristics are different from unit to unit, and when they are inputted as-is in to the differential amplifying circuits of the next stage, then the differences in the thermal characteristics between U11 and U12 will be amplified, rather than canceled, producing a temperature-based drift in the measurement value for the flow rate. Because typically the flow rate signal level is about 200 μV p-p per 1 m/s in the flow rate, this cannot be ignored.
Because of this, in the conventional technology the DC component of the outputs of U11 and U12 are cut through inserting coupling capacitors C11 and C12 into the input side of the differential amplifying circuits. However, there is a problem in that the coupling capacitors C11 and C12 have an adverse effect on the common-mode rejection ratio (hereinafter termed “CMRR”) in the differential amplifiers, and causes recovery from an anomalous input to be slow.
Adverse Effect on CMRR
The adverse effect on the CMRR in the differential amplifying circuits will be explained first.
The purpose for a differential amplifying circuit is to amplify the flow rate signal while rejecting the common-mode noise component that is superimposed on the flow rate signal. However, even though, in FIG. 6, an operational amplifier that has a high CMRR when it is by itself is used for U13, the CMRR in a differential amplifying circuit is also affected greatly by the matching of the resistive element R11 that connects between the inverting input of the differential amplifying circuit and the inverting input terminal of U13 and the resistive element R12 that connects between the non-inverting input of the differential amplifying circuit and the non-inverting input terminal of U13, the matching between the resistive element R13 that connects between the inverting input terminal and the output terminal of U13 and the resistive element R14 that connects between the non-inverting input terminal of U13 and the ground electropotential, and also the matching between the coupling capacitors C11 and C12.
Moreover, in the circuit structure in FIG. 6, the effective resistance of C11, R11, and R13, and the effective resistance of C12, R12, and R14, each form high-pass filters. As a result, if the time constants are not adequately large when compared to the magnetic excitation frequency (=signal frequency) so as to not lose the signal frequency component, the amplitude of the signal waveform will be attenuated. Because of this, it is necessary for C11 and C12 to have large capacitances of at least several tens of μF.
FIG. 8 is a signal waveform diagram illustrating the inputs and outputs of a conventional differential amplifying circuit.
For example, when the magnetic excitation frequency=12.5 Hz, R11=R12=10 kΩ, and R13=R14=100 kΩ (where the differential amplification gain is 10×) in the circuit in FIG. 6, if C11=C12=100 μF, then, as illustrated in FIG. 8(b), the flow rate signal can be amplified with essentially no smoothing of the waveform.
On the other hand, with C11=C12=10 μF, then, as in FIG. 8(a), the output voltage waveform V11 of U13 will be smoothed. In this way, any slight differences in temperature characteristics between C11 and C12 when sampling a smoothed waveform will have a large effect on the measured value for the flow rate.
Because of this, tantalum electrolytic capacitors, which have relatively good temperature characteristics, are used for the coupling capacitors C11 and C12. While multilayer ceramic capacitors (of the temperature compensating type), which have excellent capacitive precision, may be considered, but they cannot be used because they are only manufactured up to about 0.1 μF.
However, the accuracy of the capacitance of tantalum electrolytic capacitors is no more than about ±10%, at best, so mismatching between C11 and C12 is unavoidable. In particular, the impedances of C11 and C12 gets large on the low-frequency side, so the amount of the impedance mismatch becomes large as well, having an adverse effect on the CMRR of the differential amplifying circuits. Because of this, if there is a common-mode noise component in the low-frequency domain, it cannot be rejected adequately, which will cause drift in the measured value for the flow rate. Moreover, while the stability can be improved by performing a smoothing process in software for this drift, the responsiveness would suffer commensurately.
Note that if the effective resistance of R11 and R13, and the effective resistance of R12 and R14, are made large, then the impedances of C11 and C12 will be relatively small, thus making it possible to ameliorate the adverse effect of mismatch between C11 and C12 on the CMRR; however, when these values are made large, then the thermal noise due to the resistance will increase, which will have an adverse effect on the signal-to-noise ratio of the output signal. Because of this, the effective resistance must be kept to no more than about several kΩ.
Moreover, even when the signal level is adequately large so that any adverse effect on the signal-to-noise ratio due to thermal noise is not a problem, still it is necessary to maintain the matching between R11 and R12, and the matching between R13 and R14, even when the effective resistances are large, and because resistors that have both high resistance and high accuracy are not manufactured, it would be necessary to create the high resistance with high accuracy by connecting multiple low-resistance high-accuracy resistors in series, which would result in increased costs.
Slow Recovery from Anomalous Inputs
The slow recovery from anomalous inputs will be explained next.
When the fluid within the pipe goes to empty so that the detecting electrodes of the detector no longer contact the fluid, or when a signal line in the detector is broken or disconnected, or when noise beyond the tolerable input voltage range is superimposed on the input signal, or the like, flow rate measurement would become impossible, and thus the input signal fault procedure described above is performed; however, once the input signal has returned to normal thereafter, the flow rate measurement must be restarted as soon as possible.
At this time, when the gain of the differential amplifying circuit through U13, in the circuit in FIG. 6, is large, then the time for recovery from a saturated output state of U13 will be long because of the effects of the voltages with which the coupling capacitors C11 and C12 are charged, and thus the gain of U13 must not be too large. Because of this, an amplifying circuit through U14 is inserted into a subsequent stage, to divide into multiple stages, to perform amplification that enables sampling and holding at the latter stage, and to perform A/D conversion.
For example, if the pipe becomes empty and thereafter is returned to a state wherein it is filled with water, then, as illustrated in FIG. 7, if the detecting electrodes of the detector are disposed so as to be perfectly level of each other, a s shown in FIG. 7, the detecting electrode TA and the detecting electrode TB would contact the fluid simultaneously, but if disposed so as to be inclined, then the detecting electrode TB would contact the fluid first, and the detecting electrode TA would contact the fluid thereafter, producing a shift in the timing of the contact with the fluid.
FIG. 9 is a signal waveform diagram illustrating the operation of the differential amplifying circuit of FIG. 6, where (a) is when the gain is 50× and (b) is when the gain is 10×. Note that here the power supply voltage is ±5 V, and U11 through U13 are of the rail-to-rail input/output type.
In both FIGS. 9(a) and (b), in the interval prior to time mark T0, the detecting electrode TA is in a state wherein it is not in contact with the fluid (a high impedance state), and because the A terminal is lowered to the negative-side power supply voltage level by the input bias current of U11, the output of U11 will be in the saturated state on the negative side (−5 V), so the voltages VC11 on both ends of C11 will be charged to −5 V.
Moreover, if the detecting electrode TA is restored from the non-contact state to a state wherein it contacts the fluid at time mark T0, then a normal signal will be inputted into U11 from the detecting electrode TA, to return the output voltage of U11 from −5 V correctly to ±100 μV (half of the signal amplitude). At this time, the input voltage V10 of the differential amplifying circuit will be at about −5 V due to the voltage VC11 with which C11 is charged, so the output voltage V11 will be in a saturated state (−5 V) on the negative side.
While thereafter the voltage that has been charged into C11 is discharged through the series resistances of R11 and R13, for the case of the gain of 50× in FIG. 9(a), the discharging of C11 will take time commensurate with the magnitude of the resistance value of R13, and additionally, because the gain is high as well, returning from the state wherein the output is saturated to a state wherein it is possible to perform the measurements properly will take about 40 seconds.
On the other hand, in the case wherein the gain is 10×, in FIG. 9(b), the resistance value of R13 is small, so C11 is discharged quickly, and because the gain is also small, recovery from the state wherein the output is saturated to a state wherein the measurement can be performed properly will take about 10 seconds.
Consequently, because the amount of time until measurements can be performed properly becomes long when the differential amplification gain of U13 is large, it has been necessary to divide the amplifying circuit into a plurality of stages, and to insert coupling capacitors C13 and C14, as in FIG. 6, for example, at the outputs of the respective amplifiers, to cut the DC component caused by the offset voltage.
As described above, even though it is possible to return to normal operation after the input signal has been at an aberrant value in the signal amplification circuit, these capacitors C11 and C12 cause the recovery of the output signal to be slow, delaying restarting of the flow rate measurement.
The present invention is to solve such a problem, and an aspect thereof is to provide a signal amplifying circuit technology for an electromagnetic flow meter wherein it is possible to avoid a reduction in the CMRR in the differential amplifying circuit and to avoid slow recovery after an aberrant input, which are caused by the coupling capacitors.